Single and polyphase electromagnetic induction machines having regulated polar magnetic symmetry

ABSTRACT

An induction motor includes a stator having at least one pair of stator windings and a rotor with rotor windings which are magnetically coupled to the stator windings via a circumferential air gap. The rotor windings are connected together in a squirrel cage or a wound rotor configuration. The stator windings are connected in series across the source. A capacitor is connected in parallel with one of the stator windings and this combination is connected in series with the other stator winding and is sized to form a quasi-double-resonant circuit, i.e., a quasi-parallel resonant circuit with the one winding and a quasi-series resonant circuit with the other winding. The stator windings are then grouped to form definite polar areas in the stator and a balanced rotating magnetic field is produced by all the windings throughout the entire load range when the motor is connected to a power source. Both a single-phase and a polyphase motor can be configured as a quasi-double-resonant circuit with respect to each input power phase. A further polyphase motor is also described with primary stator windings connected to each power phase input and interleaved secondary stator winding magnetically coupled to the primary stator windings but not directly connected to the power inputs. The secondary stator windings have capacitors coupled in parallel thereto to form parallel floating resonant circuits. In all of these motors, the power factor is in the range of 0.96 to 1.00, but normally closer to unity due to the resonant behavior of the circuitry. The induction motor can be driven above synchronous speed to act as a generator. Also, a method of generating torque is described.

This application is a continuation-in-part of application Ser. No.900,700, filed Aug. 27, 1986.

FIELD OF THE INVENTION

The present invention relates to single and polyphase electromagneticinduction machines having regulated polar magnetic symmetry.

BACKGROUND OF THE INVENTION

With the advent of higher utility rates, power factor penalties anddemand charges, prior art induction motors have many disadvantages. Mostinduction motors in use are over-sized and inefficient. Consequentlypower bills are higher than need be due to motor inefficiency, highdemand and poor power factor (KW/KVA). As is known, the power factorinvolves the phase relationship between the a.c. voltage and the a.c.current. Utility companies generally charge a premium to the user whenthe power factor falls below 0.85 (a power factor of unity is presentwhen the voltage and current are of the same waveform completely inphase).

When energy rates were low, these drawbacks were not as important asthey now are. Often demand (the total electrical power that needs to beavailable, but not necessarily used from the line) and power factorpenalties are as much or more than the basic energy charge.

The most efficient prior art, single-phase induction motors are of thepermanent-split capacitor design, but they have low torquecharacteristics and are efficient only when the magnetic field of thedirect phase winding is balanced with that of the auxiliary phasewinding and their respective currents are displaced by 90°. In mostsplit capacitor motors, a large stator winding is directly connected tothe power terminals and a smaller auxiliary winding, serially connectedto a capacitor, is also connected across the input. The 90° displacementof current between both stator windings only exists at design load; atother load points a disproportionate distribution of magnetic fluxexists which sets up negative sequence currents in the rotor and stator,space harmonics in the air gap (e.g., the degree to which the fluxdistribution in the air gap is not sinusoidal) and high leakagereactance from the stator end turns. For example, an imbalance of phasevoltages on the order of 3% can cause a 15% to 20% increase in motorlosses.

This condition is not restricted to single-phase motors but is alsoprevalent in polyphase motors when an imbalance occurs in the polyphasevoltage supply. These losses in both single and polyphase motors candegrade insulation and reduce bearing life due to overheating of therotor and, in addition to overheating, an imbalance creates highermagnetostriction noise and poor operating performance, as can be seen inTable 1.

Another significant disadvantage is in the manufacture of new motors.Engineers are now focusing on design tolerances in an attempt toincrease motor efficiencies, producing a motor which is more susceptibleto failure due to environmental changes and bearing wear. Attempts havebeen made to create a balanced condition by a series resonating windingin combination with a phase winding but this is a tuned condition for anarrow spectrum only, and at certain load points circulating harmoniccurrents increase, and the efficiency is reduced to below that of thestandard design.

Induction motors and generators are efficient only when properly sizedto the load and when the line voltage is balanced. When operated belowdesign load or with a system imbalance, a disproportionate polarmagnetic condition exists which sets up negative sequence currents inthe rotor and stator, space harmonics in the air gap and high leakagereactance due to high currents in the phase winding. Again, an imbalancein the order of 3% can cause a 15% to 20% increase in motor or generatorlosses. This reduces insulation and bearing life and creates animbalance which is manifested as higher magnetostriction noise and pooroperating performance. Attempts have also been made to create a balancedand controlled condition in the motor by a series resonating winding incombination with a phase winding but this is a tuned condition for anarrow spectrum and at certain load points circulating harmonic currentsincrease and the efficiency is reduced to below that of the standarddesign.

SUMMARY OF THE INVENTION

The single-phase, dynamoelectric machine, which can be a motor or agenerator, includes a rotatable rotor usually in the interior spacedefined by a hollow cylindrical stationary stator. Both the rotor andstator have slots therein facing each other within which are disposedwindings. The rotor windings may be connected at each end to form asquirrel cage or brought out via slip rings. In the stator, two windingsare electrically connected in series and are circumferentially placedaround the interior surface of the hollow stator core to form magneticpoles. A capacitor is coupled in parallel with one winding and thiscombination is connected in series with the second winding. The size ofthe capacitor is such that a quasi-series resonant circuit is formedwith the second winding and a quasi-parallel resonant circuit is formedwith the first winding. The serially connected stator windings areconnected across the single-phase or polyphase power input terminals.

When power is applied to the motor, a balanced rotating magnetic fieldis generated wherein the Q factor of the circuit is continually adjustedby the admittance of the rotor windings. Because of the interactionbetween the quasi-series resonant circuit and the quasi-parallelresonant circuit, unused energy delivered to the rotor in the form ofmagnetic flux is returned via one of the stator windings and, uponcollapse of the magnetic field, the resulting voltage is stored in thecapacitor. This is due to, for example, a reduction in load torque onthe rotor. In another sense, when the torque requirements on the rotorare higher, the capacitor delivers stored energy to the appropriatewinding to compensate for the additional power requirements and maintaina balanced distribution of magnetic flux circumferentially rotatingaround the rotor.

The method of generating torque from an a.c. power source includes thestep of forming a quasi-double-resonant circuit, including a capacitiveelement which is connected in parallel to one of the inductive elementsand this combination connected in series with the other inductiveelement, providing a rotatable inductive element adapted to delivertorque; applying power across the two serially connected, stationaryinductive elements, magnetically coupling all the inductive elements andproducing a balanced rotating magnetic flux wave via the mechanismdescribed above with respect to the quasi-serial and quasi-parallelresonant circuits.

In one embodiment, the polyphase induction motor includes three pairs ofserially connected stator windings wherein a capacitor is coupled inparallel to one of the windings in each pair and the combination coupledin series with other winding in the pair to form a quasi-double-resonantcircuit. A further embodiment of the polyphase motor includes threeprimary stator windings which receive, via one of the power inputterminals, a different phase of the three-phase power applied to themotor. Three secondary stator windings are circumferentially interleavedin the stator between the three primary stator windings and aremagnetically coupled to the primary windings but are not directlyconnected to the power input terminals of the motor. A capacitor isprovided for each pair of secondary stator windings and the respectivecapacitor is in parallel with at least one secondary stator winding.Each capacitor is sized to form a quasi-parallel floating resonantcircuit with the parallel connected secondary stator winding.

Thus, it is a primary object of the present invention to eliminate orcontrol space harmonics in the air gap, negative sequence currents inthe rotor and stator windings and increase the efficiency of aninduction motor or generator.

It is another object of the present invention to increase the torquerating of a motor without increasing hysteresis loss due to magneticsaturation.

It is a further object of the present invention to improve the powerfactor of an induction motor.

It is an additional object of the present invention to store unusedenergy returned to the stator windings and deliver stored energy to themagnetic circuit upon demand.

It is still another object of the present invention to produce abalanced rotating magnetic flux wave around the rotor at substantiallyall loads.

The subject matter which is regarded as the invention together withfurther objects and advantages thereof may best be understood byreference to the following description taken in conjunction with theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagrammatical representation of a single-phase motor withregulated magnetic polar symmetry.

FIG. 2 is an electrical schematic diagram of the single-phase motor ofFIG. 1, but the circuit does not include representations of the statorcore material or the rotor material.

FIG. 3A is an oscilloscope trace of the voltage waveforms associatedwith each of the stator windings (and the capacitor) when theoscilloscope is set to trigger on the positive going slope of the supplyline voltage.

FIG. 3B is an oscilloscope trace of the current waveforms associatedwith each of the stator windings and the capacitor when the scope isalso set to trigger on the positive going slope of the supply linevoltage.

FIG. 4A is an oscilloscope trace of the line supply voltage V_(L) andthe line current I_(L) at approximately full load for a 1/4 horsepowermotor supplied with 120 volts a.c.

FIG. 4B is an oscilloscope trace of the line supply voltage and the linecurrent at approximately half-load.

FIG. 4C is an oscilloscope trace of the line supply voltage and the linecurrent at no-load.

FIG. 5 is a time-lapse illustration of an oscilloscope trace of the linesupply voltage and line current over the entire load range of the motor.

FIG. 6 is a graphic representation of the rotor current versus slipspeed in the induction motor of FIG. 1.

FIG. 7A shows the effect of resistance on the shape of a seriesresonance curve.

FIG. 7B shows the effect of L/C ratio on the shape of a series resonancecurve.

FIG. 7C shows the parallel resonance curve.

FIG. 8 is a diagrammatical representation of a polyphase induction motorwith regulated polar magnetic symmetry including a quasi-double-resonantequalizer circuit.

FIG. 9 is an electrical schematic diagram of the polyphase,quasi-double-resonant induction motor of FIG. 8 wherein the statorresonant windings are connected in a Δ configuration with respect to thesource.

FIG. 10 is an electrical schematic diagram of the polyphase,quasi-double-resonant induction motor of FIG. 8 wherein the statorresonant windings are connected in wye configuration with respect to thesource.

FIGS. 11A through 11L are diagrams showing the electric current andmagnetic conditions in a two-pole, three-phase induction motor for each30° of a complete cycle.

FIG. 12A is an oscilloscope trace of the line supply voltage V_(L) andthe line current I_(L) of one phase at full load in a 40 horsepower,three-phase, quasi-double-resonant induction motor.

FIG. 12B is an oscilloscope trace of the line supply voltage and theline current of one phase at 75% load for the motor of FIG. 12A.

FIG. 13 is a switching network for changing rotation of thedouble-resonant polyphase motor of FIG. 8.

FIG. 14A is a diagrammatical representation of a polyphase inductionmotor with regulated magnetic symmetry with a parallel floatingquasi-resonant circuit. The primary stator windings are connected in awye configuration to the source, its parallel, floating windings areconnected in a wye configuration, and the capacitors in the floatingcircuit are connected in a Δ configuration.

FIG. 14B is an electrical representation of a polyphase induction motorwith regulated polar magnetic symmetry including a parallel floatingquasi-resonant circuit with its primary stator windings connected in awye configuration to the source, its parallel floating winding are in awye configuration and the capacitors in the floating circuits are in a Δconfiguration.

FIG. 15 is an electrical diagram of the polyphase, parallel floatingquasi-resonant induction motor wherein the primary stator windings arein a wye configuration with respect to the inputs and the parallelfloating stator windings and capacitors are in a Δ configuration.

FIG. 16 is an electrical schematic diagram of a polyphase inductionmotor with regulated polar magnetic symmetry having a floatingquasi-parallel resonant design with its primary phase, stator windingsconnected in a Δ configuration with the source, its parallel floatingresonant stator windings connected in a wye configuration and thecapacitors in the floating circuits are in a Δ configuration.

FIG. 17 is an electrical schematic diagram of a polyphase, parallelfloating induction motor wherein the primary stator windings are in a Δconfiguration, the secondary stator or parallel floating windings are ina Δ configuration and the capacitors in the floating circuits are in awye configuration.

FIG. 18A is an oscilloscope trace of the line supply voltage and theline current of one phase at full load in a 40 horsepower, three-phase,quasi-parallel floating resonant induction motor.

FIG. 18B is an oscilloscope trace of the line supply voltage and theline current of one-phase at 75% load in the quasi-parallel floatingresonant induction motor.

FIG. 19 is a phaser diagram of an ideal double-resonant motor withquasi-series resonance at full-load.

FIG. 20 is the phaser diagram of a 1/3 HP motor after conversion to aquasi-double-resonant motor with quasi-series resonance at full-load.

FIG. 21 is a phaser diagram of an ideal double-resonant motor withparallel resonance at no-load.

FIG. 22 is the phaser diagram of a 1/3 HP motor after conversion to aquasi-double-resonant motor with quasi-parallel resonance at no-load.

FIG. 23 is an electrical schematic diagram of a double-resonant motorincorporating the teachings of the present invention and used inconjunction with explaining FIGS. 19 through 22.

FIG. 24 is a representation of the magnetomotive force in the air gapsurrounding the circumference of the rotor in a quasi-double-resonantmotor. Each wave represents the force (flux) in the air gap over thecircumference of the rotor at a given time in one cycle of the inputpower.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 is a diagrammatical representation of a single-phase a.c.induction motor in a squirrel cage rotor configuration. Stator ST₁ is agenerally hollow, cylindrically shaped, slotted structure of laminatedsheet steel. A rotor RO₁ is rotatably disposed in the interior space ofthe stator and is of like material. For simplification, the stator isshown as having four polar areas or teeth TA₁, TB₁, TC₁, TD₁, protrudingfrom a return magnetic path or back iron BI₁ of stator ST₁.

The actual number of poles or teeth is dependent upon physical size,horsepower and rotational speed. The physical dimensions of the motorand its integral parts are only graphically represented and theillustration does not necessarily indicate an optimum physicalconstruction. The stator or primary is shown as having two windings WAand WB commonly referred to as "wound on" teeth TA₁, TB₁, TC₁, and TD₁.As is known, the stator windings are disposed in axially extending slotson the interior of the stator. Likewise, rotor windings WC are disposedin axially extending slots on the periphery of rotor RO₁.

Stator windings WA and WB are connected in series at midpoint MP and theserial circuit is connected across power input terminals L₁ and L₂.Midpoint connection MP is passively coupled to input terminal L₁ bymeans of two capacitors CA and CB. Capacitor CB is permanently coupledwhile capacitor CA is removed from the circuit by means of centrifugalswitch CS when the motor reaches a predetermined speed during a start-upoperation. When the motor is used for low torque applications, capacitorCA and centrifugal switch CS are omitted from the circuit.

Rotor winding WC is closely magnetically coupled to stator windings WAand WB by means of the four polar areas or teeth TA₁, TB₁, TC₁, TD₁, airgap AG, the magnetic material of rotor RO₁, and the return magnetic pathor back iron BI¹.

The present invention provides a regulatory circuit that equalizes thepolar magnetic regions about the rotor regardless of the magnitude ofthe electromotive force or its wave form. This axisymmetrical alignmentof the magnetic flux reduces space harmonics in the air gap betweenstator and rotor allowing a greater net-usable flux to link the rotorwindings. This symmetry reduces the possibility of negative sequentialcurrents being established in the rotor and results in a higher torquerating for the motor without increasing hysteresis loss due to magneticsaturation. Increased efficiency is also achieved through intermediatetransfer and storage of energy and a shorter current rise time in theresonant circuit as opposed to the induction/resistance ratio in priorart induction motors.

The amount of force or mechanical torque exerted on the rotor is basedon the equation:

    F=Bl I                                                     (1)

where F is the mechanical force, B is the magnetic flux density linkingthe rotor windings, l is the physical length of the windings and I isthe current flowing in the windings. The greatest energy loss in asquirrel-cage, induction motor is heat produced as current flows throughthe windings due to the resistance of the windings.

The amount of loss is based on the formula:

    I.sup.2 R                                                  (2)

where I is the Current in amperes, and R the resistance in ohms;therefore, by increasing the net-usable flux linking the rotor (e.g.,increasing B and maintaining F and l constant in (1)), the currentcomponent I is reduced in the rotor resulting in less heat, longerbearing life and increased efficiency.

The reduction of space harmonics also reduces the eddy current effectpresent in all induction motors and generators. Since the resonantcircuit does not return unused energy to the source but rather saves itprimarily in the capacitor, it has an energy efficient topology due toits transfer of energy in phase to the rotor and transfer betweenbifilar phase pairs.

Since V_(WB) is high and I_(L) is low in the quasi-resonant circuit,core flux density is reduced and therefore hysteresis losses, eddycurrent losses and I² R losses are reduced below that of the standardmotor design. This allows the motor to operate in the linear portion ofthe BH curve. Since I_(WB) is high, the air gap flux is high andtherefore the resultant torque much higher than for the standard motor.Under heavy loads energy to the rotor can be maximized because I_(C)provides -VARS in the stator with VARS in the rotor. Maximum energytransfer to the rotor takes place when +Y(jw) rotor=-Y(jw) stator isreached. This condition is prevalent in the present invention andparticularly in the quasi-double-resonant equalizer circuit. Further,the energy transfer remains proportional throughout the entire loadrange of the motor, since the admittance of the rotor changes inproportion to its angular velocity. See FIG. 6 where the rotor currentversus slip frequency curve.

Conductance G of a resonant circuit is governed by the equation:##EQU1## where Y is the admittance (the reciprocal of impedance) oroverall ability to pass an alternating current, G equals the circuitconductance in siemens (the reciprocal of resistance) or the ability ofa pure resistance to pass electric current, B equals susceptance insiemens (the reciprocal of reactance) or the ability of inductance orcapacitance to pass alternating current B_(eq) =B_(c) -B_(L) is the netequivalent susceptance in (3). As is known, the admittance of a circuitis equal to the conductance (the real component) plus the susceptance(the imaginary component):

    Y(jw)=G(w)+jB(w)                                           (4)

At resonance the reactive power of the inductance is equal and oppositeto that of the capacitance and the source of emf to supply only thepower required by the resistance of the circuit. One importantcharacteristic of the quasi-double-resonant circuit is it check-andbalance network. FIGS. 19 through 22 contain vector diagrams for thequasi-double-resonant motor discussed in Tables 1, 2 and 3. FIG. 23 isan equivalent circuit for the motor for use in considering the vectordiagrams of FIGS. 19 through 22. It will be apparent to those skilled inthe art that at no-load the conditions exist wherein the currents in WBand CB are near equal and 180° apart, a condition referred to asparallel resonance. It will also be apparent that as the slip of themotor increases the resistance of the rot changes and, due to themagnetic coupling of the resonant windings to the rotor windings, thecircuit moves from a near parallel resonant condition to one that moreclosely resembles the conditions that exist at series resonance. Sincethe circuit still conducts current at no-load it is not at theoreticalparallel resonance and, therefore, it is referred to as quasi-parallelresonance. At full-load maximum current does not flow, or in otherwords, the current is not limited by only the resistance of the circuitand so theoretical series resonance does not exist. The circuit istherefore called a quasi-series resonant circuit. Since at both parallelresonance and series resonance the reactance of the circuit is concelledunity power factor exists. Because this condition exists in this uniquecircuit it is referred to as a quasi-double-resonant circuit. Thecircuit maintains control of both current and voltage in each windingand adjusts to maintain optimum magnetic flux conditions in each of thepolar areas. This creates a perfectly round and constant amplitude,revolving flux wave in the air gap as can be seen in FIG. 24. Thiscondition is necessary in order to allow less magnetic loss andincreased efficiency. As shown in FIG. 24, the top set of curves coversthe situation where FA is the flux in the air gap created by winding WAand the center set of curves covers the situation where FA' is the fluxcomponent created in the air gap by winding WB. The bottom curverepresents FA and FA'. As can be seen in FIGS. 19 through 24, atquasi-resonance the vector sum of the voltage drops across the capacitorCB and inductor of the series branch (winding WA) equate to line voltageand likewise the vector sum of the currents of the quasi-parallel branch(CB and WB) equate to line current. This system maintains an evendistribution of power between the two branches, as can be noted in Table3. The ratio between the reactive power of either the inductance or thecapacitance at resonance and the true power of the entire resonantcircuit is called the Q factor of the circuit. The symbol for resonantfrequency is "f_(r) ". Since at resonance, the reactance of thecapacitor equals that of the inductor (X_(L) =X_(c)), from: ##EQU2## thefollowing equation can be derived: ##EQU3## where f_(r) is in cycles persecond, L is in henrys, and C is in farads. Equation (6) shows that anRLC circuit can be brought into resonance at a certain frequency byvarying either the inductance or capacitance. It should be noted thatthe resistance of the series resonant circuit has no bearing on theresonant frequency f_(r).

The resistance governs only the Q and the minimum impedance of thecircuit at resonance and, as a result, the height of the resonance curvechanges as noted in the curves illustrated in FIG. 7A. Also in FIG. 7A,as long as the product of L and C is constant, the resonant frequency ofa series circuit is constant. However, if we increase the L/C ratio by afactor of 4:1 (see FIG. 7B), the result will be a steepening of the"skirts" of the resonance curve. Decreasing the resistance in a seriesresonant circuit and increasing the L/C ratio both have the effect ofsteepening the "skirts" of the curve by changing its height andnarrowing the resonant frequency bandwidth.

In a.c. circuits containing both inductance and capacitance, theinstantaneous energy (1/2 CV²) is stored in the capacitor as a voltageincrease, while the energy in the inductor (1/2 LI²) is stored as acurrent increase, alternately, twice each cycle. Therefore, an exchangeof reactive energy takes place between the inductance and capacitance.The source of emf (the energy supplied to the motor) is required tosupply only the difference between the reactive energy of the inductanceand the reactive energy of the capacitance. This accounts for the netreactance of a series circuit being the difference between the inductiveand capacitive reactance:

    X.sub.NET =X.sub.L -X.sub.C                                (7)

and the reactive voltage of a series circuit being the differencebetween voltage of the inductance and the voltage of the capacitance.Reactance is the imaginary component of impedance whereas resistance isthe real component thereof. This should be compared with (4).

The Q factor of a resonant circuit is the ratio of reactive power topower dissipated in the resistance. In the present case the resistanceis a representation of the mechanical load on the motor.

The resonance curve of the parallel branch or quasi-parallel resonantcircuit (CB and WB) is shown in FIG. 7C and is generally the inversefunction of the series resonant curve. Since the Q of the parallelwinding has the inverse effect on the circuit as compared with the Q ofthe series winding, the quasi-parallel resonant circuit is balanced withthe quasi-series resonant circuit and any instantaneous energy imbalancein one of those circuits is very quickly compensated for by the othercircuit. This produces the balanced rotating magnetic flux wave aroundthe rotor, as illustrated in FIG. 24.

The sensitivity of a resonant circuit can be increased by an increase inthe Q of the circuit or decreased by a decrease in its Q. Purposelyreducing the Q of a tuned circuit is called damping and when the Q=1/2,the result is called "critical damping." An example of critical dampingis the damping of the movement of a pointer in a meter to keep thepointer from oscillating. It is possible then to capitalize on thischaracteristic of resonance in the quasi-double-resonant inductionmotor. By appropriate selection of the Q factor: energy transfer to therotor can be controlled, as well as energy supplied from the source.In-rush current can be reduced since Q is the admittance magnificationfactor, and an even distribution of power from all associated motorwindings can be maintained.

The operation of a single-phase, quasi-double-resonant induction motoras shown in FIGS. 1 and 2 will now be is described. When an a.c.potential is applied to input terminals L1 and L2, capacitors CA and CBbegin to charge, capacitor CA being in the circuit due to the closure ofcentrifugal switch CS. This charging current flows through winding WAsets up a flux path which flows horizontally through teeth TA₁ and TB₁,their respective air gaps AG, rotor RO₁ and the return magnetic path orback iron BI₁. When current begins to flow in winding WA, capacitor CBbegins to charge and a potential also starts building across primarywinding WB approximately 90° out of phase with the potential across WA.See FIG. 3A where V_(WA) leads V_(WB) by 90°. Therefore, when thecurrent of WA has reached its highest magnitude, the potential across WBhas also reached its peak and current flows in winding WB.

FIGS. 3A and 3B show certain phase relationships for a single-phasemotor with regulated magnetic polar symmetry. In the example shown, themotor is a double-resonant 1/4 horsepower motor with 120 volts a.c.input.

FIG. 3A is a representation of the voltages across WA and WB,respectively. The representation clearly shows the phase relationshipsbetween V_(WA), V_(WB) and V_(CB) and this phase relationship creates abalanced rotating magnetic field. The traces in both FIGS. 3A and 3Bbegin at the positive going slope of the voltage supplied to the motor.These have shown that V_(WA) lags the supply voltage V_(L) byapproximately 45°. Turning to FIG. 3B, as the current (IW_(A)) in WAdecreases and I_(WB) builds in WB, the energy stored in the magneticfield of winding WA together with energy stored in capacitor CB, istransferred to winding WB and a rotating magnetic flux have is created.

This wave rotates from the horizontal polar axis already established inthe motor magnetic material, to a vertical position centered throughstator teeth TC₁ and TD₁, their respective air gaps, the rotor magneticmaterial, and the return magnetic path or back iron BI₁. This rotatingflux wave cuts the rotor windings WC in rotor RO¹ which causes currentto flow and consequently establishes a magnetic field in the rotor whichtries to align itself with the rotating magnetic flux wave establishedin the stator ST₁ magnetic material.

FIG. 3B is a representation of the current wave forms I_(CB), I_(WA) andI_(WB) which flow in capacitor CB and stator windings WA and WB,respectively. I_(WA) and I_(WB) display a full 90° phase shift withrespect to each other and I_(CB) is 180° out of phase with respect toI_(WB). Representations in FIGS. 3A and 3B have the same time base;therefore, a direct comparison between the voltages and currents can bemade.

When the current in WB (I_(WB)) has reached its peak it begins todecrease releasing the energy stored in its magnetic field to the rotorwinding or to capacitor CB.

Additional energy from the source through winding WA is also stored inthe capacitor CB. This pattern continues throughout each cycle ofalternating current, advancing the rotating magnetic wave one fullrevolution for each cycle. Although the illustration shows four poles,the motor is considered a two-pole motor because for every quartercycle, the magnetic field advances one quarter of a revolution.

When the motor is at rest or in a stalled condition, if voltage isapplied to input L1 and L2, the net equivalent susceptance B_(eq) ishigh since the rotating magnetic flux wave is cutting all of rotorwindings in WC at the maximum rate. This causes a considerable amount ofcurrent to flow in both stator and rotor windings, which generatemechanical torque in an attempt to bring the rotor into synchronism withthe rotating magnetic field. As the speed of the rotor increases, therate at which the windings WC are cut decreases and the net equivalentsusceptance lowers until it reaches a theoretical 0 at synchronousspeed. FIG. 6 shows the rotor current versus slip frequency curve. In alocked rotor condition, the current through the stator and rotorwindings has the same frequency as the line current. At no-load and asthe motor approaches synchronous speed, the rotor current has afrequency near 0.

Since rotor windings WC have a high coefficient of coupling to thestator windings WA and WB, any unused energy in rotor windings WC istransferred through the magnetic coupling and stored in the reactiveelements of the quasi-double-resonant equalizer circuit (WA, WB and/orCB) during oscillating load conditions. This feedback also tends tocontrol the Q factor of the resonant circuit in regulating the amount ofenergy needed from the supply (FIG. 6). Notice that at locked rotor(ORPM) the rotor current frequency is the same as line frequency, but itdecreases with speed of the rotor until at synchronous speed it toobecomes 0.

Certain facets of the resonant circuits are important. First, examineclosely the quasi-series resonant branch which consists of winding WAand capacitor CB. In FIG. 20, with full load, at first glance acondition seems to exist which violates Kirchhoff's voltage law.Measurement of the voltage drops across WA and CB, which is the same asV_(WB), determines that they are almost equal. This may infer that thetotal is double that of the source. The series resonant circuit does notviolate Kirchhoff's law since the vector sum of WA and CB equate to theinput voltage, as shown in FIGS. 19 through 22.

The ability of the quasi-series resonant circuit to produce a voltagehigher than the applied voltage is one of the most importantcharacteristics of the circuit. This is possible due to its ability tostore unused energy in WA and CB. In the series portion of the circuit,Q is the magnification (admittance) factor which determines how much thevoltage across WA and CB can increase above the applied voltage.

To separately consider the parallel branch of the quasi-double-resonantequalizer circuit which consists of WB and CB. In FIG. 22, at no-load,the current in CB leads the voltage across it by 90° and the current inWB lags its applied potential by 84.2°. Since CB and WB are parallel,the same potential appears across both (FIG. 3A) and therefore thecurrents are out of phase by 174.2°. This shows that when the current isflowing in one direction through WB, an almost equal current is flowingin the opposite direction through CB, as shown in FIG. 22.

Applying Kirchhoff's current law to midpoint MP, it is noted in FIG. 21that there is no current flowing into or out of the source at parallelresonance. The current simply oscillates back and forth between thecapacitor and winding. In the ideal parallel resonant circuit, thesource voltage would only be required to start the oscillation. Oncestarted, the source could be removed and the circuit would continue tooscillate indefinitely. Since the parallel resonant circuit displaysinverted characteristics to that of the series resonant circuit, it issometimes referred to as antiresonance. This condition exists, however,only if there are no losses in the circuit.

There are losses in the quasi-double-resonant induction motor; thegreatest loss being useful energy delivered to the load, but anotherloss to be minimized is that caused by current flowing through theresistance of the windings. Therefore, energy must continually be addedfrom the source.

This ability of a parallel resonant circuit to sustain oscillation afterthe source voltage is removed is sometimes called the "flywheel effect"and is an important feature in the motor circuit since for everyoscillation a magnetic field is built around WB. When this fieldcollapses the energy in the magnetic field induces an electromotiveforce (V) in CB. This method of energy transfer is a most efficienttopology since it is intermediate in nature and does not return unusedenergy to the source. Since the two stator windings are in series andthe two resonant circuits are basically opposites (FIGS. 7A and 7C),they tend to electrophysically control or regulate each other. The endresult is an induction motor with regulated magnetic polar symmetry overits entire range of points.

FIGS. 4A, 4B and 4C are representations of the current (line currentI_(L)) relationship with respect to the applied electromotive force(line voltage V_(L)) in a single-phase motor with regulated magneticpolar symmetry of the same type as referred to with reference to FIGS.3A and 3B. FIG. 4A is at approximately full load, FIG. 4B is atapproximately half load and FIG. 4C at no-load. It should be noted thatline current I_(L) remains closely in phase with the line voltage V_(L).Consequently, the power factor of the circuit is near unity over theentire load range. FIG. 5 is a time exposure representation of theentire load range for the single-phase motor of FIGS. 3A and 3B.

FIG. 5 shows the phase relationship of line voltage and currentthroughout the entire load range.

Another unique and desirable characteristic of the motor is theflattened current wave form I_(L). Since I_(L) is non-sinusoidal, itsRMS or effective value is considerably higher than that of a sine wavewith the same peak value and results in a magnetic field that remainshigh for a long time without saturating the iron. This maintains hightorque and reduces the hysteresis losses in the magnetic core material.The motor operates in the linear portion of the BH curve.

The present invention virtually eliminates the preponderance of currentproblems associated with conventional induction motors and generators.It is to be noted that, although the description of the invention ismade in reference to a motor, the device can operate as a generator iftorque is applied to the rotor and the device is driven abovesynchronous speed. The generator need not be supplied with reactivepower since the power factor of the device is unity. The claims aremeant to encompass this usage of the device.

The conventional polyphase motor has by design a rotating magnetic fluxwave which is unregulated and under certain operating conditions it canbecome distorted non-symmetrized.

This distortion or magnetic irregularity effects a decrease in theoperating efficiency of conventional motors. The present inventionprovides a regulatory circuit to equalize the polar magnetic regionsregardless of the magnitude of electromotive force or its wave form.This axisymmetrical alignment of the magnetic flux reduces undesiredspace harmonics in the air gap between stator and rotor allowing agreater net-usable flux to link the rotor windings as illustrated inFIG. 24.

Magnetic symmetry reduces the hazards of negative sequence currentsbeing established in the rotor and results in a higher torque valuewithout increasing hysteresis loss due to magnetic saturation. Increasedefficiency is also achieved through intermediate transfer and storage ofenergy and a shorter current rise time in the resonant circuit asopposed to that in a standard motor.

The polyphase motors are discussed below. Some important characteristicsof the quasi-double-resonant circuit are its check and balance network,its ability to produce a rotating magnetic vector, and the ability ofthe circuit to act as a phase doubler. Therefore, coil placement shouldbe so as to enhance the rotating sinusoidal magnetic wave. One exampleof proper coil placement is shown in FIG. 8, but it should be understoodthat coil arrangement and number of poles can be varied so as to producea motor having different operating characteristics. Hence, the inventionis not limited to the illustrated embodiment in FIG. 8. As with thesingle-phase motors of the present invention, at quasi-resonance thevector sum of the voltage drops across the capacitor and inductor of thequasi-series branch equate to line voltage (FIG. 19) and further thevector sum of the currents of the quasi-parallel branch equate to linecurrent (FIG. 19).

Two electrical schematics of the quasi-double-resonance polyphase motorare illustrated in FIGS. 9 and 10. The configuration shown in FIG. 9 isdiagrammatically illustrated in FIG. 8. In FIG. 8, the windings of eachquasi-resonant circuit are separated by 90° electrical. It needs to beunderstood that this angle can be adjusted to produce different torqueand operating characteristics for the motor.

FIG. 8 is a diagrammatical representation of a quasi-double-resonantpolyphase a.c. induction motor of the squirrel cage design. It has asheet-steel laminated stator ST₂ and a rotor RO₂ of like material. Forsimplification, the stator is shown as having 12 poles or teeth TA, TB,TC, etc., through and including TL protruding from a return magneticpath or back iron BI₂ ; the actual number of teeth being dependent uponphysical size, horsepower, and rotational speed. The physical dimensionsof the motor and its integral parts are for graphical representationonly and do not indicate its optimum physical construction.

The stator is shown as having three sets of quasi-double-resonantcircuits or one set per input phase. The first quasi-double-resonantcircuit consists of serially connected windings WBa and WAa are wound onteeth TA, TB, TC, and TD with the windings being connected in series atmidpoint MPa across inputs A and B.

With the windings thus connected, the midpoint is then passively coupledto input terminal A by means of capacitor CBa, i.e., in parallel towinding WBa and in series with WAa. The rotor winding WC is magneticallycoupled to the stator windings WBa and WAa by means of the four polarareas or teeth TA, TB, TC, TD, their respective air gaps AG, the rotormagnetic material RO₂ and the return magnetic path or back iron BI₂.

The second set of quasi-resonant windings WBb and WAb are connected toinput terminals B and C. They are wound on teeth TE, TF, TG and TH andare likewise connected in series at midpoint MPb, the connection ofwhich is passively coupled to input terminal B by means of capacitorCBb. The rotor winding WC is also closely coupled to the stator windingsWBb and WAb by means of the four polar areas or teeth TE, TF, TG, TH,their respective air gaps AG, the rotor magnetic material RO₂ and thereturn magnetic path or back iron BI₂.

The third set of quasi-resonant windings WBc and WAc are connected toinput terminals A and C. They are wound on teeth TI, TJ, TK and TL, andare connected in series at midpoint MPc. The midpoint connection MPc ispassively coupled to input terminal C by means of capacitor CBc. Thesecondary winding WC is closely coupled to the primary windings WBc andWAc by means on the four polar areas or teeth TI, TJ, TK, TL, therespective air gaps AG, the rotor magnetic material RO₂ and the returnmagnetic path or back iron BI₂.

The operating principles of the quasi-double-resonant polyphase motorshown in FIG. 8 are as follows. When a polyphase a.c. potential isapplied to input terminals A, B, and C, with A being 0 and goingpositive, capacitor CBa begins to charge. This charging current flowsthrough winding WAa setting up a magnetic flux path through stator teethTC and TD, their respective air gaps, the rotor magnetic material andthe return magnetic path or back iron BI₂. At the same instant of time,since three phases are acting on the stator concurrent a conditionexists similar to that shown in FIG. 11A.

FIGS. 11A through 11L illustrate current and flux paths through astandard induction motor for a full revolution in increments of 30°.These figures are presented to help explain the very complex conditionsthat exist in the motor and particularly in the quasi-double-resonantinduction motor. Although both prior art induction motors and thepresent invention develop a rotating magnetic flux wave, the flux wavein the conventional motor is not necessarily symmetrical or balancedunder all operating conditions as is the wave in the presentquasi-double-resonant motor. In FIGS. 11A through 11L, solid lines anddashed lines represent current flow through the stator windings and thedash-dot-dash lines represent the magnetic flux paths through the statorand the rotor. One of the differences in a structural sense between theconventional polyphase induction motors and the induction motorconstructed in accordance with the principles of the present inventionis that the quasi-double-resonant induction motor, for example, has awinding circumferentially interleaved between the stator windingscoupled to power input terminals A and B.

It should be noted that windings which are serially connected withrespect to the capacitor are identified by "A" in the term "WAa" whereasthe lower case letter refers to the phase of the winding. Hence,considering stator windings WAa and WBa, both these stator windings areserially connected together with respect to one another because of theterm "a," winding WAa is serially connected to capacitor CBa and statorwinding WBa is parallelly connected to capacitor CBa.

Returning to FIGS. 11A through 11L, one of the physical differencesbetween the conventional motor and the motor of the present invention isthat the present invention includes an additional winding interposedbetween the two primary stator windings. Referring to FIG. 11A, thecircumferential disposition of teeth TA, TK and TE is shown in thefigure. Referring jointly to that figure and FIG. 8, power input phase Ais applied to winding WBa wound on tooth TA at the same time thatcurrent flows through winding WBb wound on tooth TE. However, windingWAc is wound on tooth TK, as well as tooth TL, and that winding iscircumferentially interleaved between windings WBa and WBb. Thecircumferential location of the other teeth of the stator is not shownin FIG. 11A in order to simplify the drawing.

FIG. 11B illustrates that the rotor has turned 30° clockwise due to therotating magnetic flux wave. In this instance, flux waves cut statorwinding WAc wound on tooth TK as well as tooth TL.

As soon as the potential between power terminals A and B has reached itspeak, the energy stored in capacitor CBa begins to discharge intowinding WBa. This energy together with that stored in the magnetic fieldof WAa is to winding WBa setting up a magnetic flux path through teethTA and TB, their respective air gaps, the rotor return magnetic materialand the return magnetic path or back iron BI. This flux path would besimilar to that shown in FIG. 11L that illustrates in general sense themotor at 330°. A potential is also building in a positive direction withrespect to terminal C, thus causing current to flow in stator windingWAc in an attempt to charge capacitor CBc. As this current begins toflow in WAc, the magnetic field in WBc begins to collapse and the energystored in WBc together with the energy now flowing in winding WAc isstored in capacitor CBc and in the new magnetic field in WAc. This movesthe position of the magnetic flux wave 30° clockwise to that shown inFIG. 11A at 0°. This process continues as long as the polyphase sourceis applied to the motor terminals A, B and C. Consequently with each newcycle of alternating current, the windings in the motor marked with theprefix WA, i.e., WAx where x is a, b or c (the windings seriallyconnected to the capacitors), pass an electric current in an attempt tocharge their respective capacitors. Therefore, a magnetic field isestablished in these windings and an electric field or potential isdeveloped across the associated capacitors.

With the collapse of each magnetic field in the motor, the energy storedin the field is converted to an electric current which is stored eitherin its related winding pair or its associated capacitor. On the otherhand, as each capacitor discharges, the stored energy is converted to anelectric current which flows through either of its related windings.

Hence, a very efficient system for regulated transfer and exchange ofenergy is established and, at the same time, a rotating field is createdsimilar to that shown in the diagrams of FIGS. 11A through 11L. Themagnetic center of the rotating flux wave moves one tooth in theclockwise direction for every 30° advance of the polyphase source. Itcan also be seen, that a check and balance network exits. Excess energyfrom any phase in the circuit is transferred directly or indirectly toanother part of the network, since all windings have a high coefficientof coupling to each other. This balanced rotating flux wave or patterncontinues throughout each cycle of alternating current, advancing therotating magnetic flux wave one full revolution.

When the motor is at rest or in a stalled condition, if a polyphasesource is applied to motor terminals A, B and C, the net equivalentsusceptance is high since the rotating magnetic flux wave is cutting allof the rotor windings in WC. This causes a considerable amount ofcurrent to flow in both stator and rotor windings most of which isconverted to mechanical torque in an attempt to bring the rotor intosynchronism with the rotating magnetic field. As the speed of the rotorincreases, the rate at which the windings are cut and the net equivalentsusceptance lowers until it reaches a theoretical 0 at synchronousspeed. Since rotor windings WC are closely magnetically coupled to theprimary stator windings, any unused energy in rotor windings WC isreturned through the magnetic coupling and stored in the reactiveelements (WAx, WBx or CBs, where x is a, b or c) or the stator circuitryas explained earlier. This feedback also adjusts the Q factor of theresonant circuit which regulates the amount of energy needed from thesupply. At locked rotor or ORPM the frequency of the rotor current isthe same as line frequency, but as FIG. 6 shows it decreases with speedof the rotor until at synchronous speed it too becomes 0.

FIGS. 12A and 12B are representations of the relationship between theline current (I_(L)) and the applied electromotive force (line voltageV_(L)) for one phase of the three phase power line connected to thequasi-double-resonant, polyphase motor, which is a double-resonant 40horsepower motor. The power input was 460 volts three-phase. FIG. 12A isthe relationship of line voltage with respect to line current atapproximately full load, FIG. 12B is the relationship of line voltagewith respect to line current at approximately 75% load. It should benoted that the line potential V_(L) is in phase with the line currentI_(L) and that the power factor of the circuit is near unity over theentire load range as illustrated in FIG. 5.

Another unique and desirable characteristic is that of the flattenedwave form in the current component. Since I_(L) is non-sinusoidal, itsRMS or effective value is considerably higher and results in a magneticenergy transfer that is consequently more intense than that produced bya standard sine wave of current. This reduces the hysteresis loss in themagnetic core material due to the fact that the current does not drivethe core into saturation to accomplish the same work as someconventional motors, thereby reducing the coercive force or energyneeded to return the magnetic material to 0. The reduction of harmonicsalso tends to reduce eddy current losses in the magnetic core material.

In working with the quasi-double-resonant polyphase motor, it wasdiscovered that the quasi-double-resonant equalizer circuit seems tolimit the use of the polyphase motor, since the rotation of the motorcould not be changed simply by reversing the input phase sequence as iscommon with the present day standard design (A-B-C change to A-C-B).Also different operating voltages require a variation in the amount ofcapacitive reactance needed to maintain the proper Q factor.

The disadvantages can be overcome with a switching network shown in FIG.13. Switching of rotation of the motor is accomplished by two reversingcontactors (C1, C2) and (C3, C4). C1 and C3 are closed while C2 and C4remain open for one direction of rotation. The reverse switchingsituation exists for rotation in the opposite direction. The networkbasically consists of two reversing motor contactors connected such thatthe top two reverse the phase sequence and bottom two reverse thewindings with respect to the phase change.

By sacrificing the admittance magnification factor of the seriesresonance branch, similar result. i.e., a balanced rotating magneticflux is achieved by connecting the primary stator windings directly tothe input polyphase source, e.g., winding, WAa to terminal A and theneutral point forming a wye configuration as in FIG. 14B; a secondarystator winding, being connected in a Δ (FIG. 15) or wye (FIG. 14B)configuration and left floating or closely magnetically coupled to theprimary stator winding, e.g., WBa'. The angular displacement of eachwinding can be modified to give the motor different operatingcharacteristics. Hence, the invention is not limited to any set angulardisplacement of the windings.

The parallel floating concept is shown in the electrical schematic ofFIGS. 14A, 14B, 15, 16 and 17. Since the floating stator windings(secondary) are only inductively coupled to the power source, turns ofthe secondary can be chosen so as to allow the most economical amount ofcapacitive reactance to be used. The capacitor CBx, x being a, b or c,is connected in parallel with WBx so as to form a parallel resonantcircuit, the Q factor of which is determined similar to that of thequasi-double-resonant circuit.

The combination of the floating quasi-parallel resonant circuit andphase winding operate in similitude to that of the quasi-double-resonantcircuit. Energy is transferred magnetically between bifilar pairs andthe rotor winding. The source sees the primary winding as one having 0reactance or unity power factor and therefore has to supply only thepower required by the mechanical torque of the rotor and the resistanceof the circuit. The circuit allows for change in rotor rotation bysimple reversal of the incoming phase sequence. A dual voltage primarycan also be used without changing the capacitive reactance of theresonant circuit.

The quasi-double-resonant polyphase motor can be used in applicationswhich require greatly reduced inrush current control but do not need tobe reversible in their operation. For motors needing higher startingtorque and the capability of being readily reversible, the floatingparallel resonant motor is more suitable.

The secondary stator windings WBx can also be connected as though theywere individual single-phase circuits. Each of these connections givesthe motor different operating characteristics, such as that achievedwith wye- Δ starting etc.

FIG. 14A is a diagrammatical representation of a parallel-resonant orparallel floating polyphase, a.c. induction motor having a squirrel cagerotor design. The motor includes a sheet-steel laminated stator ST₂ anda rotor RO of like material. For simplification, the stator is shown ashaving twelve poles or teeth TA, TB, TC, etc., through and including TLprotruding from a return magnetic path or back iron BI₂ ; the actualnumber of teeth being dependant upon physical size, horsepower androtational speed for the motor. The physical dimensions of the motor andits integral parts are only graphically represented herein and hence theillustrations do not indicate the motor's optimum physical construction.The stator includes three primary phase windings which can be connectedto the source in a Δ or wye configuration and three sets of floatingparallel resonant circuits, one set per input phase. The three primaryphase windings WAa', WAb' and WAc' are connected to input terminals A, Band C in the wye configuration.

Three secondary stator windings WBa', WBb' and WBc' are part of thefloating parallel resonant circuits and are connected in FIG. 14 in awye configuration and in parallel with three capacitors CBa', CBb' andCBc' that are connected in a Δ configuration with respect to each other.In this circuit, capacitor CBb' is parallel to secondary stator windingsWBb' and WBc'; however, as shown in FIG. 15, the parallel floatingcapacitor need only be parallel to one secondary stator winding to formthe floating parallel resonant circuit.

The floating parallel circuitry consists of secondary windings WBa',WBb' and WBc' and capacitors CBa', CBb' and CBc'. The secondary statorwindings are wound on teeth TC, TD, TG, TH, TK and TL, respectively. Theprimary stator phase windings WAa', WAb' and WAc' are wound on teeth TA,TB, TE,TF, TI and TJ, respectively. The secondary windings arecircumferentially interleaved between the primary windings, e.g., WBa',is between WAa' and WAb'. The floating circuit is magnetically coupledto the primary phase windings and rotor RO. The actual phasedisplacement between the two primary winding sets can differ from thatshown, producing everything from a close coupling to a near unitycoupling. These variances effect desirable changes in motor operatingcharacteristics and therefore the invention is not limited to theembodiment shown in FIG. 14A.

The following is a brief description of the operating principles of themotor shown in FIG. 14A. When a polyphase a.c. potential is applied toinput terminals A, B and C, the primary phase windings, WAa', WAb' andWAc' produce a rotating magnetic flux wave similar to that shown inFIGS. 11A through 11L for the standard motor design, since they areconnected to the source in similitude to the windings of the standardmotor design. As this magnetic flux wave rotates in the stator'smagnetic material the flux cuts the floating parallel windings WBa',WBb' and WBc' together with windings WC in rotor RO. This generates apotential in the windings of the floating circuit and causes a currentto flow in them setting up a magnetic field in their associated teeth,their respective air gaps, the rotor magnetic material RO, and thereturn magnetic material or back iron BI₂. The energy stored incapacitors CBa', CBb' and CBc' is discharged into their respectivewindings in the form of an electrical current.

This ability of the quasi-parallel-resonant circuit to sustainoscillation reflects the flywheel effect and is a valuable feature ofthe motor. The parallel floating circuit not only provides the necessarymagnetizing current but the circuit tends to equalize the polar magneticenergy since it is floating in nature and thus regulates the energy flowto the windings of the rotor. This provides an intermediate exchange ortransfer of energy between bifilar windings and is consequently a mostefficient topology. Since unused energy is stored in the reactiveelements of the motor, the source need only supply the energy needed toprovide the necessary mechanical torque and of course replace anyexpended energy. The motor therefore runs at or near unity power factorthroughout its entire load range. See FIGS. 18A and 18B and note theflattened top of the current waveform I_(L). FIGS. 18A and 18B show therelationship of line voltage with respect to line current at full-loadand at 75% load, respectively, for a motor of the type discussed inconnection with FIGS. 12A and 12B.

This current waveform is very desirable since the RMS energy level ishigher than that of a sinusoidal wave resulting in more energy transferwithin the same time frame. Thus a lower current component is neededresulting in less copper losses in the associated motor windings.Another important characteristic is the symmetry of the magnetic fluxwave cutting the winding WC in the rotor RO. This symmetry orphysiomagnetic regulation created by intermediate exchange of energybetween bifilar windings results in a higher net magnetic coupling ofthe rotor and therefore reduced losses. This particular topology alsoallows for direct motor reversal without reconfiguration of the motorwindings or elaborate switching mechanisms. Since the parallel branchwindings are floating they are similar to that of the secondary of atransformer. This allows for a selection in both turns and connectionwhich will reduce the cost and physical size of the capacitors neededfor the appropriate Q factor. Although a three-phase motor is describedherein, the claims are meant to cover polyphase motors configured inaccordance with the principles of this invention.

In connection with the inventive features of the present invention, a1/3 HP Marathon single-phase induction motor was put through severaltests. Table 1 compares the performance of an original 1/3 HP Marathonsingle-phase induction motor and the same motor rewound incorporatingthe quasi-double resonant technology:

                                      TABLE 1                                     __________________________________________________________________________    1/3 HP Marathon Single-Phase Induction Motor-                                 Model:SPB56C17F5302A                                                          Marathon versus Quasi-Double-Resonant                                                    Before Conversion                                                                           After Conversion                                     Load       No Load Full Load                                                                           No Load Full                                         __________________________________________________________________________    Horsepower   0.00    0.3301                                                                              0.00    0.3303                                     Line Voltage                                                                              230.352                                                                               230.256                                                                             230.01  230.976                                     Line Amperes                                                                               2.5227                                                                                2.9492                                                                              0.5454                                                                                1.4455                                     Line Frequency                                                                            60.0    60.0  60.0    60.0                                        RPM        1792.8  1759.5                                                                              1795.2  1760.4                                       Torque (oz. in.)                                                                           0.0    11.82                                                                                0.0    11.82                                       Watts       154.7493                                                                              421.4933                                                                            122.0228                                                                              331.5053                                    Volt Amperes                                                                              581.1090                                                                              679.0709                                                                            125.4475                                                                              333.8758                                    VARS        560.1253                                                                              532.4243                                                                            29.1121                                                                               39.7151                                     Power Factor                                                                              26.63%  62.07%                                                                              97.27%  99.29%                                      Efficiency    --    58.4235%                                                                              --    74.28%                                      Ambient Temp.                                                                             22.2507                                                                               22.8080                                                                             22.4563                                                                               22.9109                                     Case Temperature                                                                          50.7649                                                                               51.1410                                                                             39.4897                                                                               39.7817                                     Bearing Temp.                                                                             45.0604                                                                               46.5713                                                                             36.3931                                                                               37.9134                                     Breakdown Torque                                                                          35.58 (oz. in.)                                                                             35.58 (oz. in.)                                     Locked Rotor Torque                                                                       37.62 (oz. in.)                                                                             54.42 (oz. in.)                                     Locked Rotor                                                                              15.37          8.301                                              Amperes                                                                       Sound Pressure                                                                Level      85 dB SPL     81 dB SPL                                                       (Linear)      (Linear)                                             __________________________________________________________________________

Table 2 gives a comparison of the winding specifications of the originalMarathon motor with those of the quasi-double resonant motor.

                  TABLE 2                                                         ______________________________________                                        1/3 HP Marathon Single-Phase Induction Motor -                                Model: SPB56C17F5302A                                                         Span   Cord Factor   Turns   Effective Turns                                  ______________________________________                                        Marathon versus Quasi-Double-                                                 Resonant Winding Specifications                                               Marathon Start Winding                                                        1-8    .9807853      37      36.28906                                         1-6    .8314693      33      27.43850                                         1-4    .5555703      29      16.11154                                         Total Effective Turns/Pole                                                                          79.83910                                                Wire Size: 1#20 AWG                                                           Pole Displacement 90 Electrical Degrees                                       Marathon Run Winding                                                          1-8    .9807853      68      66.69340                                         1-6    .8314693      52      43.23642                                         1-4    .5555703      32      17.77825                                         Total Effective Turns/Pole                                                                         127.70810                                                Wire Size: 1#19 AWG                                                           Pole Displacement 90 Electrical Degrees                                       Quasi-Double-Resonant Winding Specifications                                  Double-Resonant Winding WA                                                    1-8    .9807853      50      49.03927                                         1-6    .8314693      54      44.89936                                         1-4    .5555703      32      17.77825                                         Total Effective Turns/Pole                                                                         111.71690                                                Wire Size: 1#23 and 1#24                                                      Pole Displacement 90 Electrical Degrees                                       Double-Resonant Winding WB                                                    1-8    .9807853      50      49.03927                                         1-6    .8314693      54      44.89936                                         1-4    .5555703      32      17.77825                                         Total Effective Turns/Pole                                                                         111.71690                                                Wire Size: 1#22 and 1# 23                                                     Pole Displacement 90 Electrical Degrees                                       ______________________________________                                    

Table 3 gives test data on the Marathon motor after conversion of themotor to quasi-double-resonant circuitry.

                  TABLE 3                                                         ______________________________________                                        1/3 HP Marathon Single-Phase Induction Motor-                                 Model: SPB56C17F5302A                                                         Data After Conversion To Quasi-Double-Resonant Circuitry                                                Power                                                         Voltage                                                                              Amperes  Factor    Watts                                     ______________________________________                                        No-Load                                                                       Winding WA  153.40   0.5458   72.87%  61.0109                                 Winding WB  188.20   2.5760   12.50%  60.6004                                 Capacitor CB (43                                                                          188.20   3.0100    0.0009%                                                                              0.5098                                  mfd)                                                                          Power Supply                                                                              230.00   0.5458   97.27%  122.1069                                Full-Load                                                                     Winding WA  158.10   1.4410   69.00%  157.1973                                Winding WB  167.10   1.9660   53.00%  174.0107                                Capacitor CB (43                                                                          167.00   2.6570    0.0009%                                                                              0.3993                                  mfd)                                                                          Power Supply                                                                              230.00   1.4410   99.9%   331.0986                                Locked Rotor                                                                  Winding WA  111.00   8.310    77.17%  711.8238                                Winding WB  153.10   10.960   70.19%  1177.7714                               Capacitor CB (43                                                                          153.10   2.422     .0003% .1112                                   mfd)                                                                          Capacitor CA (145                                                                         153.10   8.550     .0008% 1.0472                                  mfd)                                                                          Power Supply                                                                              230.00   8.310    98.93%  1890.8491                               ______________________________________                                    

All test procedures were conducted in compliance with IEEE standard112-B. In the tests, Lebow torque sensors and Ohio Semitronics, Inc.watt transducers, voltage transducers and current transducers were allcalibrated with test systems that are within current calibrationrequirements traceable to the U.S. National Bureau of Standards. AMagtrol hysteresis brake was used to load the motor under test. In allcases, an appropriately sized Lebow in-line rotating shaft torque sensorwas connected between the motor and the load. Signals were processedthrough a Daytronic strain gauge conditioner and fed to a Compudascomputer Also fed to the computer was information from the electronicprecision watt transducers, RMS voltage transducers, RMS currenttransducers, frequency transducers and thermocouples. The computercontinuously integrated the data from all of these sources and compiledit into coherent form for collection, processing, display andcommunication. Also part of the test system was a large variabletransformer which allowed testing with balanced voltage or with anydesired degree of imbalanced voltage.

All readings were taken after the motor was run in a loaded conditionfor 30 minutes. Sound pressure levels were obtained at a distance of onefoot in a Ray Proof double wall sound room by a Quest Electronics Model215 Sound Level Meter.

As shown in Table 2, with the motor in original manufacturers conditionthe effective turns of the two windings in the motor are not equal(start winding 79.8 turns and run winding 127.7). After remanufacture toquasi-double-resonant condition, the two windings have equal turns (eachhave 111.7). This modification allows the creation of a perfectly roundrevolving magnetic field. The quasi-double-resonant technology usessmaller diameter wire to increase the packing factor and allow for themanufacture of smaller motors and/or motors with smaller magneticlosses. As can be seen from the above figures, the wire utilized is lessthan half the size of the original wire.

Table 3 shows that the power factor stays near unity throughout theentire load range (97.27 at no-load to 99.9 at full-load). Because ofthe accuracy of the test equipment, it is easily seen that the total ofthe watt measurements for each component equals the amount of wattagetaken from the power supply. The energy (watts) in the two windings isnearly equal, at both no-load and full-load, even though the voltage,current and power factor are not equal.

Finally, Table 1 shows that, after the conversion toquasi-double-resonant circuitry, the motor has less slip, higherefficiency, higher power factor, reduced temperature, lower sound level,reduced in-rush current and increased starting torque.

The claims appended hereto are meant to cover modifications disclosedand undisclosed which come within the scope of the claims.

What is claimed is:
 1. An electromagnetic induction machine comprising:arotatable, magnetically excitable rotor; a stationary stator operativelyassociated with said rotor; at least two windings electrically connectedin series and each winding defining a pair of magnetic poles disposedproximate said rotor in said stator; and capacitor means in parallelcombination with one of said windings and said combination in serieswith the other one of said windings, the size of the capacitor beingsuch that a quasi-parallel resonant circuit is formed with said onewinding and a quasi-series resonant circuit is formed with said otherone of said windings.
 2. An induction machine as claimed in claim 1,wherein said machine has at least one pair of power input terminalswhich are adapted to be coupled to an a.c. power source and the seriallyconnected windings are coupled across said input terminals.
 3. Aninduction machine as claimed in claim 2, wherein said rotor includes aplurality of longitudinally insulated conductors, said statorcircumferentially surrounds said rotor and includes teeth radiallyextending towards said rotor, and said windings establish said magneticpoles via said teeth when excited, wherein a balanced rotating magneticfield is present due to the transfer of energy from the excited windingsto said rotor, the return of energy from said rotor to said windings andthe storage of the returned energy in said capacitor.
 4. Anelectromagnetic induction machine as claimed in claim 1, wherein saidrotor circumferentially surrounds said stationary stator.
 5. Anelectromagnetic induction machine as claimed in claim 1, wherein saidcapacitor means comprises at least one alternating-current bipolar,non-electrolytic liquid-type capacitor.
 6. An electromagnetic inductionmachine as claimed in claim 1, further comprising a second capacitormeans arranged in series with a switch means, said arrangement being inparallel with said capacitor means, said switch means being normallyclosed, and opened when said rotor reaches a certain speed.
 7. Anelectromagnetic induction machine as claimed in claim 6, wherein saidsecond capacitor means being selected from the group consistingessentially of a.c. bipolar, non-electrolytic liquid-type capacitors anda.c. bipolar electrolytic-type capacitors.
 8. An electromagneticinduction machine as claimed in claim 2, wherein said capacitor means insaid two quasi-resonant circuits serve as a phase doubling capacitor bycreating two balanced phases from said input power source.
 9. Anelectromagnetic induction machine as claimed in claim 1, wherein thephase voltages generated for each of said two windings is approximatelyequal to the applied phase voltage divided by the square root of two.10. An electromagnetic induction machine as claimed in claim 1, whereinsaid serially connected windings are wound with effective turns rangingfrom the same number of turns in each winding to a ratio of 1.05:1. 11.An electromagnetic induction machine as claimed in claim 1, wherein saidserially connected windings substantially are wound with wire sizeranging from the same size in each winding to a ratio of 1:2.
 12. Anelectromagnetic induction machine as claimed in claim 11, wherein theserially connected windings, when wound with unequal wire sizes, havingthe quasi-series winding as the winding with the smaller circular milarea and the quasi-parallel winding as the winding with the largercircular mil area.
 13. An electromagnetic induction machine as claimedin claim 1, further comprising an air gap between said teeth and saidrotor, wherein said air gap includes a perfectly round revolvingmagnetic field.
 14. An electromagnetic induction machine as claimed inclaim 1, wherein due to the reduction of current in the windings, saidmachine is operative in the linear portion of the BH curve belowsaturation.
 15. An electromagnetic induction machine as claimed in claim1, wherein the quasi-resonant windings are placed relative to each otherin the range between 60 and 130 electrical degrees.
 16. Anelectromagnetic induction machine as claimed in claim 15, wherein thequasi-resonant windings are placed at substantially 90 electricaldegrees relative to each other.
 17. An inductive dynamoelectric machinecomprising:a hollow, cylindrically shaped, longitudinally slottedstator; at least a pair of stator windings disposed in the slots aroundthe interior of said stator, said stator windings being electricallyserially connected together and connected across a power input of saiddynamoelectric machine; a capacitor in parallel combination with one ofsaid stator windings and said combination in series with the other saidstator winding forming a quasi-double-resonant circuit that includes aquasi-series resonant circuit and a quasi-parallel resonant circuit; alongitudinally slotted, rotatable rotor disposed in the interior spacedefined by said stator; and a plurality of electrically coupled rotorwindings disposed in the slots on the periphery of said rotor whereinsaid stator windings and said rotor windings are magnetically coupledtogether and a balanced rotating magnetic flux wave is produced due tothe storage and delivery of energy by the capacitor, the stator windingsand the rotor windings under substantially all load conditions.
 18. Aninductive dynamoelectric machine as claimed in claim 17, wherein thecapacitance value of said capacitor is selected to produce saidquasi-double-resonance circuit and wherein the Q factor of saidquasi-double-resonance circuit is continually adjusted by the admittanceof the rotor windings.
 19. An electromagnetic induction machine asclaimed in claim 17, wherein said capacitor means comprises at least onealternating-current bipolar, non-electrolytic liquid-type capacitor. 20.An electromagnetic induction machine as claimed in claim 17, furthercomprising second capacitor means arranged in series with a switchmeans, said arrangement being in parallel with said capacitor means,said switch means being normally closed, and opened when said rotorreaches a certain speed.
 21. An electromagnetic induction machine asclaimed in claim 20, wherein all second capacitor means being selectedfrom the group consisting essentially of a.c. bipolar, non-electrolyticliquid-type and a.c. bipolar, electrolytic-type capacitor.
 22. Anelectromagnetic induction machine as claimed in claim 17, wherein saidcapacitor means in said two quasi-resonant circuits serve as a phasedoubling capacitor by creating two balanced phases from said input powersource.
 23. An electromagnetic induction machine as claimed in claim 17,wherein the phase voltages generated for each of said windings areapproximately equal to the applied phase voltage divided by the squareroot of two.
 24. An electromagnetic induction machine as claimed inclaim 17, wherein said serially connected windings are wound witheffective turns ranging from the same number of turns in each winding toa ratio of 1.05:1.
 25. An electromagnetic induction machine as claimedin claim 17, wherein said serially connected windings substantially arewound with wire size ranging from the same size in each winding to aratio of 1:2.
 26. An electromagnetic induction machine as claimed inclaim 17, wherein the serially connected windings, when wound withunequal wire size, have the quasi-series winding as the winding with thesmaller circular mil area and the quasi-parallel winding as the windingwith the largest circular mil area.
 27. An electromagnetic inductionmachine as claimed in claim 17, further comprising an air gap betweensaid teeth and said rotor, wherein said air gap includes a perfectlyround revolving magnetic field.
 28. An electromagnetic induction machineas claimed in claim 17, wherein due to the reduction of current in thewindings, said machine is operative in the linear portion of the BHcurve or well below saturation.
 29. An electromagnetic induction machineas claimed in claim 17, wherein the quasi-resonant windings are placedrelative to each other in the range between 60 and 130 electricaldegrees.
 30. An electromagnetic induction machine as claimed in claim29, wherein the quasi-resonant windings are placed at substantially 90electrical degrees relative to each other.
 31. A polyphase inductivedynamoelectric machine adapted to be supplied with a polyphase power ata like number of power phase input terminals comprising:a rotatablerotor carrying a plurality of interconnected rotor windings; a statorcircumferentially surrounding said rotor; a pair of serially connectedstator windings for each phase of said polyphase power disposed in saidstator, each pair receiving, via one of said power input terminals, adifferent phase of said polyphase power and adapted to be magneticallycoupled to said rotor windings; and a respective capacitor for each pairof stator windings, said respective capacitor connected in series with afirst winding of said pair and in parallel with a second windingthereof, the size of the capacitor being such that a respectivequasi-series resonant circuit is formed with said first winding and arespective quasi-parallel resonant circuit is formed with said secondwinding.
 32. A polyphase inductive dynamoelectric machine as claimed inclaim 31, wherein all the pairs of stator windings are connected in a Δconfiguration with respect to said input terminals.
 33. A polyphaseinductive dynamoelectric machine as claimed in claim 31, wherein all thepairs of stator windings are connected in a wye configuration withrespect to said input terminals.
 34. A method of generating torque forman a.c. power source including the steps of:forming aquasi-souble-resonant circuit comprising two stationary seriallyconnected inductive elements and a capacitive element connected inparallel combination with one of said inductive elements and thecombination connected in series with the other of said inductiveelements to respectively form a quasi-parallel resonant circuit and aquasi-series resonant circuit; providing a rotatable inductive elementadapted to deliver torque; applying said a.c. power across said twoserially connected inductive elements; magnetically coupling said twoserially connected inductive elements with said rotatable inductiveelement; and producing a balanced rotating magnetic flux wave by storingand by delivering stored energy from one of said quasi-serial resonantcircuit or said quasi-parallel resonant circuit to the other one of saidquasi-serial resonant circuit or said quasi-parallel resonant circuitupon a change in the magnetic flux linking said two inductive elementsand said rotating inductive element.
 35. A method of generating torquefrom an a.c. power source including the steps of:forming aquasi-double-resonant circuit comprising two stationary seriallyconnected inductive elements and a capacitive element connected inseries with one of said inductive elements and in parallel to the otherof said inductive elements to respectively form a quasi-series resonantcircuit and a quasi-parallel resonant circuit; providing a rotatableinductive element adapted to deliver torque; applying said a.c. poweracross said two serially connected inductive elements; magneticallycoupling said two serially connected inductive elements with saidrotatable inductive element; and producing a balanced rotating magneticflux wave by storing and by delivering stored energy from one of saidquasi-serial resonant circuit or said quasi-parallel resonant circuit tothe other one of said quasi-serial resonant circuit or saidquasi-parallel resonant circuit upon a change in the magnetic fluxlinking said two inductive elements and said rotating inductive element.36. A polyphase inductive dynamoelectric machine adapted to be suppliedwith a polyphase power at a like number of power phase input terminalscomprising:a rotatable rotor carrying a plurality of interconnectedrotor windings; said rotor circumferentially surrounding a stationarystator; a pair of serially connected stator windings for each phase ofsaid polyphase power disposed in said stator, each pair receiving, viaone of said power input terminals, a different phase of said polyphasepower and adapted to be magnetically coupled to said rotor windings; anda respective capacitor for each pair of stator windings, said respectivecapacitor connected in quasiseries with a first winding of said pair andin parallel with a second winding thereof, the size of the capacitorbeing such that a respective quasi-series resonant circuit is formedwith said first winding and a respective quasi-parallel resonant circuitis formed with said second winding.
 37. A polyphase inductivedynamoelectric machine as claimed in claim 36 wherein all the pairs ofstator windings are connected in a Δ configuration with respect to said,input terminals.
 38. A polyphase inductive dynamoelectric machine asclaimed in claim 36, wherein all the pairs of stator windings areconnected in a wye configuration with respect to said input terminals.